Method of stepwise voltage control for supplying an induction motor

ABSTRACT

A method for stepwise voltage control using thyristors for supplying an induction motor at a fixed frequency and a variable voltage. A time sequence of electromotive force amplitudes corresponding to a predetermined law for torque variation during transient operating conditions is stored. Voltages applied to the motor and the current flow are determined to approximate the electromotive force generated by the motor. Static switch conduction intervals are used to adapt the electromotive force to the stored amplitude during the transient operating conditions.

This application is a 371 of PCT/FR95/00817 filed on Jun. 20, 1995.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention concerns a method of stepwise voltage controlusing thyristors for supplying an induction motor at a fixed frequencyand at a variable voltage, more particularly in the situation in whichthe load that it drives has a quadratic torque-speed law, noteworthy inthat it combines great simplicity of the equipment with a level ofperformance in terms of progressive stopping and starting that hashitherto been unattainable using more complex equipments such asfrequency converters.

2. Discussion of the Background

Electronic starters for induction motors have achieved significantpenetration of the industrial market in the last ten years, replacingolder generation electrotechnical hardware such as star-delta switches,autotransformers and liquid rheostats. Their success is related toadvances in power semiconductors (increased reliability and economiccompetitivity), on the one hand, and to advances in digital controlcircuits having an increasing processing capacity, on the other hand.Among the various electronic speed variation systems, the electronicstarter represents the simplest means of controlling the speed ofinduction motors: in a system of this kind, the power circuit orthree-phase stepwise voltage controller typically comprises a set ofthree alternating current switches each in series with one phase betweenthe AC line voltage and the motor, each switch typically comprising twothyristors connected in anti-parallel. The fact that these switcheschange state at a low frequency, namely that of the AC line voltage, anddo not require any turn-off control, as turn-off occurs naturally whenthe current passes through zero, also simplifies the control circuits.However, these starters can only control the amplitude of the motorvoltage, at the fixed frequency of the AC line voltage. By comparison,frequency converters have a more complex structure, as much from thepoint of view of the power circuit, typically comprising a three-phasepower transistor bridge with the rupture capacity needed to switch themotor current at a frequency of at least a few kilohertz, as from thatof the control circuit, which has to generate variable voltage andfrequency waves by pulse width modulation. To compensate this, they havea second control quantity, namely the frequency, which can be variedindependently of the magnetic flux and the torque to optimize theoperation of the motor at each point, in particular with regard to slipand losses. As a result, applications are divided between frequencyconverters and electronic starters, the former offering high performanceand the latter moderate cost.

Given the trend for increasingly higher performance of digital controlcircuits, it has now become possible to expand the field of applicationsof electronic starters into that of frequency converters. For example,starting and stopping pumps represents a particular problem due to theexistence of mechanical resonance that is manifested on the occasion ofa rapid variation in flowrate by oscillation of the fluid in the pipe,known as "water hammer". This phenomenon is harmful as much through itsreduction of the service life of the installation as through theaccompanying acoustic noise. This noise is particularly unacceptable inwater distribution installations in urban areas. Previous means ofsolving this problem tend to eliminate all sudden variation in the flow,and therefore in the speed, during stopping or starting. They use twotechniques:

modulation of the flowrate by a progressive action solenoid valve;

variation of the speed of the pump by a speed regulator, generallyconsisting of a frequency converter and an alternating current motor.

Both methods have the same drawback, namely high cost, increasingly sowith increasing power levels, the latter being routinely between 10 kWand 500 kW. Efforts to date to eliminate "water hammer" when stopping apump, by regular deceleration commanded by a basic electronic starter,have failed. It is well known that varying the voltage at constantfrequency introduces a discontinuity into the voltage-speedcharacteristic of the induction motor: below a certain speed whichadditionally depends on the speed-torque characteristic of the load butwhich, in the case of a pump, can be as high as two-thirds the nominalspeed, operation becomes unstable and during deceleration the motor"stalls" and suddenly drops to a low speed, while during accelerationthe motor "runs away" and is out of control between a low speed and thestable operating speed. This behavior causes "water hammer" and theelectronic starter has therefore proved to be unsuitable for solvingthis problem.

SUMMARY OF THE INVENTION

The aim of the present invention is to propose an electronic startercontrol method that assures stable behavior of the starter over all ofthe range of speeds between zero and the nominal speed and which, in thepresence of a load with a quadratic torque-speed characteristic, iscapable of starting and slowing performance comparable with that of afrequency converter and eliminates the phenomenon of "water hammer"during slowing of a pump, under more advantageous economic conditions.

The method of the invention is intended for a stepwise voltage controltype power circuit such as that made up of a three-phase system of threestatic switches each of which comprises two thyristors connected inanti-parallel, placed between the AC line voltage and the inductionmotor supplied with power, each in series with one phase of the AC linevoltage, or possibly of the motor alone in the case of a delta connectedmotor. It uses a phase variation thyristor control system, known initself, for example from "Induction machine SCR voltage reduction;optimized control and dynamic modelling", by A. P. Van den Bossche andJ. A. Melkebeeke, IEE Conference Publication Number 234, London 1984.Note however, that the method of varying the motor voltage describedtherein, by adjusting the angle of non-conduction of the switches, is tobe understood as constituting only one example and can be replaced byany other method, such as the more conventional method in which thecontrol magnitude is the phase at which the switches are turned onrelative to the phase of their supply voltage.

The invention therefore proposes a method for time control of transientoperating conditions of a multiphase induction motor driving a load theresisting torque of which varies with the speed in accordance with aknown law and supplied at variable voltage via static switches havingperiods of conduction of variable duration, characterized in that,having memorized a time succession of rotor electromotive forceamplitudes corresponding to a predetermined law of torque variationunder transient operating conditions, the voltages applied to the motorand the current passing through it are determined, the amplitude of therotor electromotive force developed by the motor is at leastapproximately deduced and the conduction intervals of the staticswitches are varied to adjust the amplitude of the rotor electromotiveforce developed to the corresponding amplitude memorized under thetransient operating conditions.

In accordance with an advantageous development of the invention, thevoltages applied to the motor and the current flowing through it aresampled in a substantially synchronous manner and the sampled values areconverted into digital signals, the rotor electromotive force amplitudebeing deduced from the aforementioned digital signals by computation inthe digital domain, the rotor electromotive force being treated as avector.

This approach exploits the advantageous performance of commerciallyavailable digital control circuits.

In accordance with another aspect of the invention, the set point valueof the rotor electromotive force or its approximate expression isdetermined by a time law adapted to produce the required acceleration ordeceleration. This law is based on the torque-speed characteristic ofthe driven load, and possibly on its inertia, by application of thefollowing formula:

    e=(c/g).sup.1/2

where

e is the rotor electromotive force,

g is the slip,

c is the motor torque,

e, g and c being expressed as a proportion of the nominal rotorelectromotive force, slip and torque of the motor, respectively.

The features and advantages of the invention will emerge from thefollowing description given by way of example with reference to theaccompanying drawings in which FIGS. 1 and 2 show two complementaryparts of a block diagram showing the application of the invention tocontrolling a three-phase induction motor.

In the selected embodiment shown, a short-circuit rotor three-phaseinduction motor M has three stator windings connected in a starconfiguration and supplied with power via terminals S1, S2 and S3. Thelatter are connected to the conductors of a three-phase AC line voltageV1, V2, V3 through respective switches I1, I2, I3 each made up of a pairof thyristors connected in anti-parallel. The triggers of the thyristorsare driven by the output of respective pulse transformers T1, T2 and T3.Current transformers TI1, TI2 and TI3 are connected in series with thethree phases.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a set A of elements for determining the rotor motorelectromotive force and a set B for producing the rotor electromotiveforce set point.

FIG. 2 shows a control system C for the switches I1, I2, I3.

DISCUSSION OF THE PREFERRED EMBODIMENTS

The three subassemblies are described in turn hereinafter, noting thatthey include digital processing elements although for clarity these arerepresented by means of hardwired logic symbols.

A) Measuring the Electromotive Force

Three differential amplifiers 11, 12, 13 each associated with arespective attenuator network A1, A2, A3 with an attenuationapproximately equal to 100 and each made up of four resistors, deliverthree voltages W1, W2, W3 imaging the three voltages between phases ofthe motor M1. Likewise the amplifier 10 supplies a voltage I imaging therectified value of the three-phase current from M provided by thesecondary currents of TI1, TI2 and TI3 flowing through the three-phaserectifier bridge 14 and a load resistor 15. The four signals I, W1, W2and W3 are sampled by four sample and hold circuits 16, 17, 18 and 19,respectively, and passed to an analog-digital converter 20. The samplingtime for each of the three switches I1, I2 or I3, near the middle of anon-conduction interval, is defined by the OR logic function 30, asexplained further below. Dividing them by the nominal value of thevoltage between phases Wn at 21, 22 and 23 reduces the three voltagesW1, W2, W3 to the values w1, w2, w3. Dividing it by the nominal value Inof the current from M (24) reduces I to the value i. The rotorelectromotive force e is calculated (25) from w1, w2, w3 and i using theequation:

    e={1/3 1/3(wj+1-wj-1).sup.2 +(wj-0.02 i).sup.2 !}.sup.1/2

B) Production of the Electromotive Force Set Point e ref

Consider first deceleration to a stop within a time Ta. The value of eref programmed in a memory 31 at the address n is written:

    e ref(n)=0.2(1-n/m)(n/m).sup.-1/2

with:

    0<n≦m

and:

    e≦1

m being the total numbers of registers used in the memory 31 to definethe law in accordance with which e ref varies during stopping. The stoptime Ta is varied by means of a binary switch C1, the set value c1 ofwhich initializes the counter 32 on each passage through zero. Thiscounter therefore divides the frequency f of an internal clock CLK towhich it is connected by a switch 33 by c1. In each period of thefrequency obtained in this way, the address of the memory 31 isincremented by one unit via the OR logic function 34. The theoreticalstopping time Ta is given by:

    Ta=m(c1/f)

At the start of the stopping process, the value of n is initializedduring a phase having a duration of several tens of milliseconds,defined by a monostable function 35 activated by a pushbutton 35a.During this phase, the switch 33 routes the frequency f to a switch 36which increments via the OR function 34 or decrements the address n ofthe memory, depending on whether the sign of the error

    e.sub.0 =e ref(n)-e

is positive or negative at the output of the summing device 37 whichcontrols the switch 36 using a comparator 38. The error signal e₀ isadditionally applied to a proportional-integral regulator 40 shunted bya switch 39 which prevents it acting during the time period defined bythe monostable 35. From the initial value n0 obtained, n is incrementedduring stopping by one unit, at a period of c1/f, up to the maximalvalue m.

The equation (2) given above for deceleration is justified as follows inthe case considered here of a quadradic load.

Starting from the equation e=(c/g)^(1/2) with e, c and g having therelative values already mentioned above, consider the speed w of themotor, also relative to the nominal value of that speed. This relativespeed value is therefore between 0 and 1, like the other relative valuese, c and g already considered.

The quadratic nature of the load in question (pump) implies:

    c=w.sup.2

whence

    e=w/g.sup.1/2

Consider now the nominal slip a of the motor, the value of which isroutinely around 0.04.

The relation between w and g is written

    w(1-a)=1-ag

where:

    g= 1-w(1-a)!/a

whence:

    e=a.sup.1/2 w 1-w(1-a)!.sup.-1/2

Taking a =0.04 and taking a to be negligible compared to 1:

    e=0.2 w(1-w).sup.-1/2

For starting at constant acceleration it is possible to write w=t, thetime t being expressed relative to the start time.

The following then applies:

    e=0.2 t(1-t).sup.-1/2

For stopping at constant deceleration it is possible to write:

    w=1-t

t being relative to the stopping time, whence:

    e=0.2(1-t)t.sup.-1/2

which justifies the equation (2) with:

    n/m=t

C) Thyristor Control (FIG. 2)

The proportional-integral action regulator 40 that receives as input theerror e₀ delivers a control signal d that defines the angle ofnon-conduction of the switches Ij for each half-period of the AC linevoltage. The extinction of Ij is observed with the aid of one of threecircuits made up of an attenuator network A11, A12, A13, a doublecomparator B11, B12, B13 and an OR circuit C11, C12, C13. Duringconduction the two outputs of the double comparator are low because ofthe effect of the -P bias at its input, which introduces a thresholdcorresponding to a few tens of volts at the terminals of the switches.Extinction is manifested in the appearance of a voltage at the terminalsof the switch which, as soon as it crosses the threshold imposed by thebias -P, causes one or other output of the double comparator and that ofthe OR circuit to change to the high state, depending on its sign. Themonostable D1, D2 or D3 generates a short pulse at extinction thatinitializes two groups of programmable monostables F1, F2, F3 and G1,G2, G3. The time-delay of the first group is equal to the value definedby the angle d/2 supplied by the regulator 40 and a divider 41. At theend of this time-delay the monostable F1, F2 or F3 goes high and themonostable H1, H2 or H3 generates a short pulse which triggers thesample and hold circuits 16 through 19 via the OR function 30. Thetime-delay of the second group is equal to the value defined by theangle d defined by the regulator 40. The time-delay signals delivered inthe form of short pulses by monostables K1, K2 or K3 when G1, G2 or G3goes high are applied to the primary of T1, T2 or T3 and cause thecorresponding switches to conduct. As each switch is commanded in thisway on each half-period of the AC line voltage, the stepwise voltagecontroller delivers the required voltage to the motor.

Of course, the invention is not limited to the example described, butencompasses all variant executions thereof within the scope of theclaims.

I claim:
 1. A method for controlling a multiphase induction motor during the time interval of predetermined transient operating conditions, said motor driving a load the resisting torque of which varies with the speed in accordance with a predetermined law, wherein the method comprises the steps of:supplying said motor at constant frequency and variable voltage via static switches (I₁, I₂, I₃) by varying periods of conduction of said switches; reading from a memory, in a predetermined sequence, a succession of prerecorded information representative of a succession of desired electromotive force amplitudes corresponding to a predetermined torque variation sequence during said predetermined transient operating conditions; determining, during the same said time interval, the voltages (W₁, W₂, W₃) applied to said motor and the current (1) passing through its stator; deducing from said voltages and said current, during the same time interval, the variation of the rotor electromotive force amplitude (e) developed by motor M; and controlling said static switches (I₁, I₂, I₃) by acting on their conduction intervals in order to permanently adjust said rotor electromotive force amplitude (e) to simultaneously read-in-memory electromotive force amplitude (e_(ref)) corresponding to said predetermined torque variation sequence during said predetermined transient operating conditions.
 2. Method according to claim 1 wherein said variable voltage applied to the motor and the current flowing through it are sampled substantially synchronously and the sampled values are converted into digital signals, the amplitude of the rotor electromotive force being deduced from the aforementioned digital signals by computation in the digital domain, the electromotive force being treated as a vector.
 3. The method according to claim 2 wherein the sampled values are substantially synchronized to the middle of an interval between the two conduction periods of said static switches.
 4. The method according to claim 3 wherein the sampling is synchronized to each interval between two conduction periods.
 5. The method according to claim 1 wherein the time succession of rotor electromotive force amplitudes is determined by application of the following equation:

    e=(c/g).sup.1/2

where e is the electromotive force, g is the slip, c is the motor torque, e, g and c being expressed in values relative to the nominal rotor electromotive force, slip and torque of the motor, respectively.
 6. The method according to claim 5 for a motor driving a load the resisting torque of which is a substantially quadratic function of the speed, wherein the time succession of the electromotive force amplitudes is determined by application of the following equation:

    e=w(g).sup.-1/2

where w is the speed of the motor relative to the nominal speed of the motor.
 7. The method according to claim 6 wherein the acceleration is substantially constant under transient operating conditions, and wherein the time succession of the rotor electromotive force amplitudes is determined by application of the following equation:

    e=a.sup.1/2 (1-t).sup.α t.sup.β

with

    0<t<1 and e≦1

where t represents the time relative to the duration of the transient operating conditions, a represents the nominal slip, α and β are exponents with respective values -1/2 and 1 for starting and respectively 1 and -1/2 for stopping. 